There are several power converter topologies that have been developed over the years, which are intended to improve the power density and switching efficiency of power converters. An emerging focus of new converter topologies is to provide a means to reduce or eliminate converter switching losses, while increasing the switching frequencies. Lower loss and higher switching frequency means more efficient converters, which can reduce the size and weight of converter components. Additionally, with the introduction of high speed composite semiconductor switches, such as metal oxide semiconductor field effect transistor (MOSFET) switches operated by pulse width modulation (PWM), recent forward and flyback topologies are now capable of operation at greatly increased switching frequencies, such as, for example, up to 1.0 MHz.
However, an increase in switching frequency can cause a corresponding increase in switching and component stress related losses, as well as increased electromagnetic interference (EMI), noise, and switching commutation problems, due to the rapid ON/OFF switching of the semiconductor switches at high voltage and/or high current levels. Moreover, modern electronic components are expected to perform multiple functions, in a small space, efficiently, and with few undesirable side effects. For instance, a modern voltage converter that provides for relatively high power density and high switching frequencies, should also include uncluttered circuit topologies, provide for isolation of the output or “load” voltage from the input or “source” voltage, and also provide for variable step-up or step-down voltage transformation.
FIG. 1 illustrates a conventional flyback type voltage converter. The converter 10 includes a transistor T1, a controller 14, a transformer 12, a capacitor C1, and a diode D1. Input voltage to the circuit may be unregulated DC voltage derived from an AC supply after rectification and filtering. The transistor T1 is a fast-switching device, such as a MOSFET, the switching of which is controlled by a fast dynamic controller 14 to maintain a desired output voltage Vout. The secondary winding voltage is rectified and filtered using the diode D1 and the capacitor C1. The transformer 12 of the flyback converter functions differently than a typical transformer. Under load, the primary and secondary windings of a typical transformer conduct simultaneously. However, in the flyback converter, the primary and secondary windings of the transformer do not carry current simultaneously. In operation, when the transistor T1 is turned ON, the primary winding of the transformer 12 is connected to the input supply voltage such that the input supply voltage appears across the primary winding, resulting in an increase of magnetic flux in the transformer 12 and the primary winding current rises linearly. However, with the transistor T1 turned ON, the diode D1 is reverse biased and there is no current through the secondary winding. Even though the secondary winding does not conduct current while the transistor T1 is turned ON, the load, represented as resistor Rload, coupled to the capacitor C1 receives uninterrupted current due to previously stored charge on the capacitor.
When the transistor T1 is turned OFF, the primary winding current path is broken and the voltage polarities across the primary and secondary windings reverse, making the diode D1 forward biased. As such, the primary winding current is interrupted but the secondary winding begins conducting current thereby transferring energy from the magnetic field of the transformer to the output of the converter. This energy transfer includes charging the capacitor C1 and delivery energy to the load. If the OFF period of the transistor T1 is sufficiently long, the secondary current has sufficient time to decay to zero and the magnetic field energy stored in the transformer 12 is completely dissipated.
The flyback topology has long been attractive because of its relative simplicity when compared with other topologies used in low power application. The flyback “transformer” serves the dual purpose of providing energy storage as well as converter isolation, theoretically minimizing the magnetic component count when compared with, for example, the forward converter. A drawback to use of the flyback is the relatively high voltage and current stress suffered by the switching components. Additionally, high turn-off voltage (caused by the parasitic oscillation between transformer leakage inductance and switch capacitance) seen by the primary switch traditionally requires the use of a resistor, capacitor, diode subcircuit, such as a snubber circuit. This parasitic oscillation is extremely rich in harmonics and pollutes the environment with EMI, and causes high switching losses from the switching components in the form of extra thermal dissipation.
FIG. 2 illustrates a conventional forward type voltage converter. The converter 20 includes a transistor T1, a controller 24, a transformer 22, a capacitor C1, diodes D1 and D2, and an inductor L1. As with the flyback converter, input voltage to the circuit may be unregulated DC voltage derived from an AC supply after rectification and filtering. The transistor T1 is a fast-switching device, such as a MOSFET, the switching of which is controlled by a fast dynamic controller 24 to maintain a desired output voltage Vout. The secondary winding voltage is rectified and filtered using the diode D1 and the capacitor C1. The load, represented as resistor Rload, is coupled across the rectified output of the secondary winding. The transformer 22 is desired to be an ideal transformer with no leakages, zero magnetizing current, and no losses. In operation, when the transistor T1 is turned ON, the primary winding of the transformer 22 is connected to the input supply voltage such that the input supply voltage appears across the primary winding and simultaneously a scaled voltage appears across the secondary winding. The diode D1 is forward biased when the transistor T1 is turned ON, and the scaled voltage across the secondary winding is applied to the low pass filter circuit preceding the load. The diode D2 is reverse biased and therefore does not conduct current when the transistor T1 is turned ON. In the case of an ideal transformer, no energy is stored in the transformer, unlike the flyback converter. The scaled voltage is supplied as a constant output voltage when the transistor T1 is turned ON.
When the transistor T1 is turned OFF, the primary winding current path is broken and the voltage polarities across the primary and secondary windings reverse, making the diode D1 reversed biased and the diode D2 forward biased. The result is zero current flow through both the primary and secondary windings. However, the forward biased diode D2 provides a freewheeling path for uninterrupted current to continue to flow through the inductor L1 and the load. The inductor L1 provides the magnetic flux to maintain this current flow while the transistor T1 is turned OFF. When the transistor T1 is turned OFF, there is no power flow from the input source to the load, but the output voltage is maintained nearly constant by a relatively large capacitor C1. The charged capacitor C1 and the inductor L1 provide continuity in load voltage. However, since there is no input power when the transistor T1 is turned OFF, the stored energy in the capacitor C1 and the inductor L1 slowly dissipate. The switching frequency of the transistor T1 is set to maintain the output voltage within a required tolerance.
As with the flyback converter, the non-ideal nature of the forward converter results in noise and loses that reduce efficiency.
In an effort to reduce or eliminate the switching losses and reduce EMI noise the use of “resonant” or “soft” switching techniques has been increasingly employed in the art. The application of resonant switching techniques to conventional power converter topologies offers many advantages for high density, and high frequency, to reduce or eliminate switching stress and reduce EMI. Resonant switching techniques generally include an inductor-capacitor (LC) subcircuit in series with a semiconductor switch which, when turned ON, creates a resonating subcircuit within the converter. Further, timing the ON/OFF control cycles of the resonant switch to correspond with particular voltage and current conditions across respective converter components during the switching cycle allows for switching under zero voltage and/or zero current conditions. Zero voltage switching (ZVS) and/or zero current switching (ZCS) inherently reduces or eliminates many frequency related switching losses.
The application of such resonant switching techniques to conventional power converter topologies offers many advantages for high density, high frequency converters, such as quasi sinusoidal current waveforms, reduced or eliminated switching stresses on the electrical components of the converter, reduced frequency dependent losses, and/or reduced EMI. However, energy losses incurred during control of zero voltage switching and/or zero current switching, and losses incurred during driving, and controlling the resonance means, are still problematic.
Several power converter topologies have been developed utilizing resonant switching techniques, for example U.S. Pat. No. 7,764,515 entitled “Two Terminals Quasi Resonant Tank Circuit,” to Jansen et al. (Jansen), which is hereby incorporated in its entirety by reference. Jansen is directed to a flyback type converter including a quasi-resonant tank circuit. FIG. 3 illustrates the flyback type converter of Jansen. The quasi-resonant flyback converter 30 is similar to the flyback converter 10 of FIG. 1 with the addition of a quasi resonant tank circuit formed by a transistor T2, diodes D2, D3, and D4, and capacitors C2 and C3. When the transistor T1 is turned ON, the transistor T2 is turned OFF, and the primary winding of the transformer 32 is connected to the input supply voltage such that the input supply voltage appears across the primary winding, resulting in an increase of magnetic flux in the transformer 32 and the primary winding current rises linearly. No current flows through the secondary winding of the transformer 32 because the diode D1 is reverse biased. When the transistor T1 is turned OFF, the transistor T2 turns ON parametrically, without control of a separate control circuit. The diodes D2, D3, and D4 and the capacitor C3 function as driving circuitry for the transistor T2. With the transistor T2 turned ON, the capacitor C2 is essentially coupled in parallel to the transformer 32, and the energy previously stored in the primary winding causes current to circulate in the circuit formed by the capacitor C2 and the primary winding, forming a resonant tank. As with the flyback converter of FIG. 1, energy stored in the primary winding is delivered to the load while the transistor T1 is turned OFF. However, in the quasi-resonant flyback converter 30 of FIG. 3, a portion of the resonant energy generated in the resonant tank is also delivered to the load while the transistor T1 is turned OFF and the transistor T2 is turned ON. In this manner, the quasi-resonant flyback converter 30 of FIG. 3 delivers peak energy equal to energy from the typical flyback operation plus the resonant energy. However, current flow within the resonant tank cycles between positive and negative current flow through the primary winding. The configuration of the secondary side circuit, in particular the diode D1, only allows delivery of resonant energy during one direction of primary winding current flow. Resonant energy corresponding to the other direction of primary current flow is not delivered.
In addition to providing an increase in peak energy, the quasi-resonant flyback converter of FIG. 3 provides the conventional advantages associated with a resonant circuit, such as reduced frequency dependent losses and EMI. However, the increased energy delivered by the quasi-resonant flyback converter is still less than the energy delivered by the forward converter.